Method and apparatus for symbol independent discriminator correlator automatic frequency control

ABSTRACT

A symbol independent automatic frequency controller ( 100 ) includes a symbol correlator ( 110 ) for receiving a complex digital signal generated within a selective call receiver wherein the symbol correlator provides a real signal ( 114 ), an imaginary signal ( 116 ), and a magnitude signal ( 112 ), a discriminator ( 140 ) coupled to the symbol correlator for receiving and processing the real signal and the imaginary signal to provide a frequency error signal, a low pass filter ( 120 ) and comparison element ( 130 ) for filtering the magnitude signal and comparing the magnitude signal to a predetermined lock threshold signal that controls the bandwidth of the symbol correlator in the event the magnitude signal exceeds the lock threshold signal, and a feedback loop providing the imaginary signal ( 116 ) to an accumulator ( 170 ) and back to the symbol correlator wherein the accumulator provides the symbol correlator with an apriori-symbol phase feedback ( 172 ).

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. application Ser. No.09/359,553, filed Jul. 22, 1999 now U.S. Pat. No. 6,493,406, andassigned to Motorola, Inc.

FIELD OF THE INVENTION

The present invention is directed to a method and apparatus in acommunication device, such as a selective call receiver, and moreparticularly to a communication device capable of eliminating a symbolphase and frequency difference between transmitted symbols and apriorisymbols generated within a correlator.

BACKGROUND OF THE INVENTION

The Maximum Likelihood Detector which is also known as the OptimumNoncoherent Detector (or correlation detector) for detecting frequencyshifted keyed (FSK) signals in an additive white gaussian noise channelis well known. The performance of a correlation detector can also beachieved with other detector architectures such as a Matched Filter or aFast Fourier Transform (FFT). However, the ability of these detectors toachieve a significant sensitivity improvement (i.e., 4 dB) over that ofdiscriminators for M-level orthogonal signaling (i.e. FLEX®) or up to 3dB improvement for 4-level quasi-orthogonal signaling (i.e., ReFLEX®)depends greatly on the frequency offset between the transmitter andreceiver. For example, acceptable performance of the correlationdetector (in a FLEX receiver) requires the transmitter's carrierfrequency and receiver's local oscillator frequency to match better thanabout 0.2 parts per million (ppm) at 900 MegaHertz. Stable frequencyreferences accurate to this level of precision are not available at areasonable cost. At least 5 to 10 ppm of residual frequency offset erroris typical in today's radios. Thus, what is needed is a correlationdetector having an automatic frequency control (AFC) system capable ofproviding acceptable performance in terms of eliminating frequencyoffset error at very low cost.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an AFC mixed-signal feedback loop inaccordance with the present invention.

FIG. 2 is a more detailed block diagram of an AFC mixed-signal feedbackloop using complex signals in accordance with the present invention.

FIG. 3 is a block diagram of an AFC circuit in accordance with thepresent invention.

FIG. 4 is a block diagram of a symbol correlator used in the AFC circuitin accordance with the present invention.

DESCRIPTION OF A PREFERRED EMBODIMENT

FIG. 1 illustrates a basic AFC mixed-signal feedback loop 10 inaccordance with the present invention showing a mixer 30 mixing a radiofrequency signal 20 with a reference frequency signal provided by avoltage controlled oscillator (VCO) 90. The mixer 30 provides a receivedsignal to an Automatic Frequency Controlled element 100 or AFC corewhich provides an output to control the VCO 90. The VCO 90 in turn feedsback the reference frequency to the mixer. As will be shown in FIG. 3,the AFC element 100 uses a symbol correlator and discriminator toprovide a method, independent of transmitted symbol, of eliminatingfrequency offset in a selective call receiver between a frequencytransmitted to the selective call receiver and a reference frequency atthe selective call receiver. In summary, this method preferably includesthe basic steps of receiving the radio frequency signal 20 into themixer 30 which in turn provides a received signal to the AFC element100. The output of the AFC element 100 controls a VCO 90 wherein the VCO90 in turn feeds back a reference frequency to the mixer 30.

The AFC element 100 uses a novel symbol-correlator architectureutilizing apriori-symbol phase-feedback to analyze the mixer 30 output.The resulting output of this correlator subsystem, consists mainly of asingle frequency component representative of instantaneous frequencyoffset, independent of transmitted FSK symbol, and with substantiallyimproved signal to noise ratio. The apriori-symbol phase-feedbackeliminates the average symbol to symbol phase changes which wouldotherwise occur in the complex correlator output. As the AFC loopsettles, the apriori-symbol phase-feedback provides for a smoother andmore continuous complex output from the correlator as the transmittedsymbols change. The control signal output from this correlator is thenfurther analyzed within AFC element 100 by a discriminator and finallythen averaged by an accumulator. The output from this accumulatorcontrols the frequency of the FracN Synthesizer (or VCO) and eliminatesthe frequency offset in a selective call receiver between a frequencytransmitted to the selective call receiver and a reference frequency atthe selective call receiver. The apriori-symbols within AFC element 100serve as the reference frequency within the receiver.

Referring to FIG. 2, a more detailed block diagram of an AFCmixed-signal feedback loop 10 using complex signals in accordance withthe present invention is shown. As in FIG. 1, a radio frequency signal20 is mixed at a mixer 30 with a local oscillator signal or referencesignal. The reference signal is preferably generated by a fractional Nsynthesizer and/or VCO 90. Additionally, the reference signal should bea complex signal having in-phase and quadrature portions. Likewise, themixer output provides a complex signal analog input to a post mixeramplifier and filtering element 40 for amplifying and filtering. Theelement 40 provides a complex analog input signal to a sigma-deltaanalog to digital converter element 50. The element 50 converts ananalog signal and provides a complex digital signal to another element60 that performs the functions of decimating, channel low pass filteringand direct current offset correction. The output of element 60 providesa baseband output (in-phase output portion 62 and quadrature outputportion 64) to a demodulator (not shown). This same output is still acomplex digital signal which serves as an input signal to a limiterelement 70. The output of element 60 may actually be multiple paralleloutputs, but the limiter element 70 provides a single output for each ofthe in-phase and quadrature portions of the complex signal designated asthe LIM I (72) and LIM Q (74) signals respectively. Signals 72 and 74provide a complex input signal to the AFC element 100. The output of theAFC element 100 provides a control input 82 to the fractional Nsynthesizer and/or VCO element 90. The reference trim register 85 can beutilized during manufacture or servicing to trim out a large crystaloscillator to within acceptable tolerance levels. This allows for a lessexpensive crystal to be used. The AFC reference trim register 85 can beadded to the control input 82 using a summing element 80 to provide anadjusted control input 84 to the element 90. The register 85 valuesimply offsets F_(OUT) by a fixed value, and has no affect on thedynamic behaviour of the Discriminator Correlator AFC loop over itsfrequency ‘pull-in’ range.

The AFC element 100 also provides an AFC Lock signal 101 which is fedback to a symbol correlator within AFC element 100 for bandwidth controlas shown in FIG. 3.

FIG. 3 is a block diagram showing the AFC element 100 in greater detail.The AFC element serves as a symbol independent automatic frequencycontroller having a symbol correlator 110 for receiving a complexdigital signal (signals 72 and 74) generated within a selective callreceiver wherein the symbol correlator 110 provides a real output signal114, an imaginary output signal 115, an imaginary signal 116 and amagnitude signal 112. A discriminator 140 is preferably coupled to thesymbol correlator 110 for receiving and processing the real signal 114and the imaginary signal 115 to provide a frequency offset error signal141. The frequency offset error signal 141 is accumulated in anaccumulator 150 and further processes the output of the discriminator140. The accumulated signal maybe limited by upper and lower thresholds(upper and lower rail limits) to set a maximum frequency offset range ofwhich the AFC element 100 will control. The output from the accumulator150 provides the control input 82 to the Fraction N synthesizer and/orVCO 90 (see FIG. 2).

A low pass filter 120 filters the magnitude signal and a comparisonelement 130 compares the magnitude signal to a predetermined lockthreshold value that controls the bandwidth of the symbol correlator 110in the event the magnitude signal exceeds the lock threshold value. Thiscircuitry serves to narrow the AFC element's symbol-correlator bandwidthwhen the residual frequency-offset is sufficiently small. The resultantnarrow correlator bandwidth further improves the signal to noise ratioof the control signal fed to the discriminator, and therefore allows forfiner control of residual frequency offset error.

The AFC element 100 further comprises a feedback loop providing theimaginary signal 116 to an accumulator 170 and back to the symbolcorrelator 110 wherein the accumulator 170 provides the symbolcorrelator with an apriori-symbol phase feedback 172. The imaginarysignal 116 is preferably an apriori symbol phase error signal.Preferably, the imaginary signal 116 is accumulated, scaled andnegatively fedback to correct the apriori-symbols' phase to assist innulling out the average intersymbol phase discontinuities that mayexist. The AFC element in this manner also operates to rapidly eliminatefrequency offset on strong or weak signals at sensitivity levelsindependent of a transmitted set of data.

Referring to FIG. 4, a block diagram is shown of the symbol correlator110 used in the AFC element 100 in accordance with the presentinvention. The limited complex rectangular input signal (Lim I+j*Lim Q(74,72)) is preferably multiplied by complex +/−4800 Hz apriori signals(201) in complex multipliers 200 and 300. Preferably, dual-bandwidthrectangular window filters 202 and 302 filter the outputs from thecomplex multipliers 200 and 300 respectively. These window filtersworking as low-pass filters exhibit a sin(x)/x magnitude response. In awide-band mode, filters 202 and 302 preferably have a 5-samplerectangular impulse response resulting in a sin(x)/x magnitude responsewith a corresponding first null at 9600 Hz (the sampling rate is 48kHz). In a narrow-band mode, filters 202 and 302 preferably have a15-sample rectangular impulse response resulting in a sin(x)/x magnituderesponse with corresponding first null at 3200 Hz (again, the samplingrate is 48 kHz).

The wide-band mode for filters 202 and 302 is selected during initialfrequency acquisition to ideally meet a +/−5 kHz frequency ‘pull-in’range. The AFC_LOCK signal 101 (see FIGS. 3 and 4) preferably becomesactive when the residual frequency offset is less than about 2 kHz anddynamically selects the narrow-band mode for filters 202 and 302. Thisresults in a significant improvement in signal to noise ratio, allowingfor accurate fine-frequency offset acquisition on noisy signals.

The Imaginary (IM) output components from filters 202 and 302 aremultiplexed through switches 208 and 308 respectively. The signal withthe greatest energy is selected at a node that provides imaginary signal116. The imaginary signal 116 is accumulated (170), scaled (171) andnegatively fed back to correct the apriori-symbols' phase (201) assignal 172. The apriori-symbol phase-feedback 172 thus nulls averageintersymbol phase discontinuities. The resulting complex signal RE_OUT,IM_OUT (signal 113) is primarily a continuous-phase single-frequencycomplex tone of frequency equal to the instantaneous frequency offset,independent of received symbol.

This complex rectangular correlator signal 113, (RE_OUT+j*IM_OUT) isthen operated on by the discriminator 140. The discriminator 140 is acomplex discriminator whose output signal 141 corresponds to a valueproportional to instantaneous frequency of its complex input.Mathematically, the discriminator 140 operation is equivalent to:OUT141=RE_OUT*D(IM_OUT)−IM_OUT*D(RE_OUT), where the operator ‘D(IM_OUT)’corresponds to the time-derivative of IM_OUT and the operator‘D(RE_OUT)’ corresponds to the time-derivative of RE_OUT.

The output signal 141 from the discriminator is accumulated inaccumulator 150 to produce the frequency control signal F_(OUT) (82), asdepicted in FIG. 3. F_(OUT) (82) dynamically controls the FRAC N SYNTH &VCO block 90 as depicted in FIG. 2. The Discriminator Correlator AFCloop (FIG. 2), utilizes the properties of negative feedback toadaptively null the frequency offset at the complex baseband Output(nodes or signals 62 and 64).

Note, in FIG. 4, the letter “C” designates a complex signal. Note thatthe magnitude signal 112 is provided conventionally by summing (252) thesquares of the magnitudes (204 or 304) of the complex signals from thepost correlator filters 202 and 302 respectively. Note also that theappropriate branches for signals 113 and 116 with the greatest magnitudeare selected by the comparison logic element 250 via switches 206 or 306for signal 113 or via switches 208 or 308 for signal 116.

In another aspect of the present invention, a method is shown in FIG. 4of determining a carrier frequency offset in a correlator independent ofdata modulating a carrier frequency. The method comprises the steps ofcreating a local apriori-symbol phase signal from an imaginary portionof a received complex signal within the correlator and eliminating anaverage symbol to symbol phase difference between the received complexsignal and the local apriori-symbol phase signal. Then a series ofindividual complex correlator components are summed to provide a singletone output representative of the carrier frequency offset. The methodmay further comprise the step of providing a control signal output to afractional N synthesizer wherein the control signal output is a feedbackcontrol signal developed by accumulating a discriminated correlatorsignal.

The above description is intended by way of example only and is notintended to limit the present invention in any way except as set forthin the following claims.

What is claimed is:
 1. A symbol correlator for receiving a complexdigital signal and providing a real output signal, an imaginary outputsignal and an imaginary signal, comprising: complex multipliers formultiplying apriori symbol signals; dual bandwidth filters coupled tothe complex multipliers for filtering corresponding outputs from thecomplex multipliers and for providing corresponding outputs that aremultiplexed for negatively feeding back the imaginary signal to correctan apriori's symbols' phase; magnitude squared elements for squaring themagnitude of the real output signal and the imaginary output signal fromthe corresponding dual bandwidth filters for receiving a complex outputand providing a continuous-phase single-frequency complex tone offrequency equal to an instantaneous frequency offset.
 2. The symbolcorrelator of claim 1, wherein the imaginary signal is accumulated andscaled before being negatively fed back.
 3. The symbol correlator ofclaim 1, wherein limited complex signals are multiplied by complexapriori signals in the complex multipliers.
 4. The symbol correlator ofclaim 1, wherein the dual bandwidth filters operate in both a wide-bandmode and a narrow-band mode.